Scheme of a switching laboratory power supply based on TL494. Switching laboratory power supply for TL494 Switching power supply diagram for TL494

This project is one of the longest I have done. One person ordered a power supply for a power amplifier.
Previously, I had never had the opportunity to make such powerful pulse generators of a stabilized type, although I have experience in assembling IIP quite big. There were many problems during assembly. Initially, I want to say that the scheme is often found on the Internet, or more precisely, on the website, an interval, but.... the scheme is initially not ideal, has errors and most likely will not work if you assemble it exactly according to the scheme from the site.

In particular, I changed the generator connection diagram and took the diagram from the datasheet. I redid the power supply unit of the control circuit, instead of parallel-connected 2-watt resistors, I used a separate 15 Volt 2 Ampere SMPS, which made it possible to get rid of a lot of hassle.
I replaced some components to suit my convenience and launched everything in parts, configuring each node separately.
A few words about the design of the power supply. This is a powerful switching network power supply based on a bridge topology, has output voltage stabilization, short-circuit and overload protection, all these functions are adjustable.
The power in my case is 2000 watts, but the circuit can easily remove up to 4000 watts if you replace the keys, the bridge and fill it with 4000 uF of electrolytes. Regarding electrolytes, the capacity is selected based on the calculation of 1 watt - 1 µF.
Diode bridge - 30 Ampere 1000 Volt - ready-made assembly, has its own separate airflow (cooler)
Mains fuse 25-30 Ampere.
Transistors - IRFP460, try to select transistors with a voltage of 450-700 Volts, with the lowest gate capacitance and the lowest resistance of the open channel of the switch. In my case, these keys were the only option, although in a bridge circuit they can provide the given power. They are installed on a common heat sink; they must be isolated from each other; the heat sink requires intensive cooling.
Soft Start Mode Relay - 30 Amp with 12 Volt Coil. Initially, when the unit is connected to a 220 Volt network, the starting current is so high that it can burn the bridge and much more, so a soft start mode is necessary for power supplies of this rank. When connected to the network through a limiting resistor (a chain of series-connected resistors 3x22Ohm 5 Watt in my case), the electrolytes are charged. When the voltage on them is high enough, the control circuit power supply (15 Volt 2 Ampere) is activated, which closes the relay and through the latter the main (power) power is supplied to the circuit.
Transformer - in my case, on 4 rings 45x28x8 2000NM, the core is not critical and everything connected with it will have to be calculated using specialized programs, the same with output chokes of group stabilization.

My unit has 3 windings, all of them provide bipolar voltage. The first (main, power) winding is +/-45 Volts with a current of 20 Amps - for powering the main output stages (current amplifier) ​​of the UMZCH, the second +/-55 Volts 1.5 Amps - for powering the diff stages of the amplifier, the third +/- 15 for powering the filter unit.

The generator is built on TL494, tuned to 80 kHz, beyond the driver IR2110 to manage keys.
The current transformer is wound on a 2000NM 20x12x6 ring - the secondary winding is wound with 0.3mm MGTF wire and consists of 2x45 turns.
In the output part, everything is standard; a bridge of KD2997 diodes is used as a rectifier for the main power winding - with a current of 30 amperes. The bridge for the 55 volt winding is UF5408 diodes, and for the low-power 15 volt winding - UF4007. Use only fast or ultra-fast diodes, although you can use regular pulse diodes with a reverse voltage of at least 150-200 Volts (the voltage and current of the diodes depends on the winding parameters).
The capacitors after the rectifier cost 100 Volts (with a margin), the capacity is 1000 μF, but of course there will be more on the amplifier board itself.

Troubleshooting the initial circuit.
I will not give my diagram, since it is not much different from the one indicated. I will only say that in circuit 15 we unhook the TL pin from 16 and solder it to pins 13/14. Next, we remove resistors R16/19/20/22 2 watts, and power the control unit with a separate power supply of 16-18 Volts 1-2 amperes.
We replace resistor R29 with 6.8-10 kOhm. We exclude the SA3/SA4 buttons from the circuit (under no circumstances short them! There will be a boom!). We replace R8/R9 - they will burn out the first time they are connected, so we replace them with a 5-watt 47-68 Ohm resistor; you can use several series-connected resistors with the specified power.
R42 - replace it with a zener diode with the required stabilization voltage. I highly recommend using all variable resistors in the circuit of the multi-turn type for the most accurate settings.
The minimum limit for voltage stabilization is 18-25 Volts, then the generation will fail.

SWITCH POWER SUPPLY FOR TL494 AND IR2110

Most automotive and network voltage converters are based on a specialized TL494 controller, and since it is the main one, it would be unfair not to briefly talk about the principle of its operation.
The TL494 controller is a plastic DIP16 package (there are also options in a planar package, but it is not used in these designs). The functional diagram of the controller is shown in Fig. 1.

Figure 1 - Block diagram of the TL494 chip.

As can be seen from the figure, the TL494 microcircuit has very developed control circuits, which makes it possible to build converters on its basis to suit almost any requirements, but first a few words about the functional units of the controller.
ION circuits and protection against undervoltage. The circuit turns on when the power reaches the threshold of 5.5..7.0 V (typical value 6.4V). Until this moment, the internal control buses prohibit the operation of the generator and the logical part of the circuit. The no-load current at supply voltage +15V (output transistors are disabled) is no more than 10 mA. ION +5V (+4.75..+5.25 V, output stabilization no worse than +/- 25mV) provides a flowing current of up to 10 mA. The ION can only be boosted using an NPN emitter follower (see TI pp. 19-20), but the voltage at the output of such a “stabilizer” will greatly depend on the load current.
Generator generates a sawtooth voltage of 0..+3.0V (the amplitude is set by the ION) on the timing capacitor Ct (pin 5) for the TL494 Texas Instruments and 0...+2.8V for the TL494 Motorola (what can we expect from others?), respectively, for TI F =1.0/(RtCt), for Motorola F=1.1/(RtCt).
Allowable operating frequencies from 1 to 300 kHz, with the recommended range Rt = 1...500 kOhm, Ct = 470pF...10 μF. In this case, the typical temperature drift of frequency is (naturally, without taking into account the drift of attached components) +/-3%, and the frequency drift depending on the supply voltage is within 0.1% over the entire permissible range.
For remote shutdown generator, you can use an external key to short-circuit the Rt input (6) to the ION output, or short-circuit Ct to ground. Of course, the leakage resistance of the open switch must be taken into account when selecting Rt, Ct.
Rest phase control input (duty factor) through the rest phase comparator sets the required minimum pause between pulses in the arms of the circuit. This is necessary both to prevent through current in the power stages outside the IC, and for stable operation of the trigger - the switching time of the digital part of the TL494 is 200 ns. The output signal is enabled when the saw exceeds the voltage at control input 4 (DT) by Ct. At clock frequencies up to 150 kHz with zero control voltage, the resting phase = 3% of the period (equivalent bias of the control signal 100..120 mV), at high frequencies the built-in correction expands the resting phase to 200..300 ns.
Using the DT input circuit, you can set a fixed rest phase (R-R divider), soft start mode (R-C), remote shutdown (key), and also use DT as a linear control input. The input circuit is assembled using PNP transistors, so the input current (up to 1.0 μA) flows out of the IC rather than into it. The current is quite large, so high-resistance resistors (no more than 100 kOhm) should be avoided. See TI, page 23 for an example of surge protection using a TL430 (431) 3-lead zener diode.
Error Amplifiers - in fact, operational amplifiers with Ku = 70..95 dB at constant voltage (60 dB for early series), Ku = 1 at 350 kHz. The input circuits are assembled using PNP transistors, so the input current (up to 1.0 μA) flows out of the IC rather than into it. The current is quite large for the op-amp, the bias voltage is also high (up to 10 mV), so high-resistance resistors in the control circuits (no more than 100 kOhm) should be avoided. But thanks to the use of pnp inputs, the input voltage range is from -0.3V to Vsupply-2V
When using an RC frequency-dependent OS, you should remember that the output of the amplifiers is actually single-ended (series diode!), so it will charge the capacitance (upward) and will take a long time to discharge downward. The voltage at this output is within 0..+3.5V (slightly more than the generator swing), then the voltage coefficient drops sharply and at approximately 4.5V at the output the amplifiers are saturated. Likewise, low-resistance resistors in the amplifier output circuit (feedback loop) should be avoided.
Amplifiers are not designed to operate within one clock cycle of the operating frequency. With a signal propagation delay inside the amplifier of 400 ns, they are too slow for this, and the trigger control logic does not allow it (side pulses would appear at the output). In real PN circuits, the cutoff frequency of the OS circuit is selected on the order of 200-10000 Hz.
Trigger and output control logic - With a supply voltage of at least 7V, if the saw voltage at the generator is greater than at the DT control input, and if the saw voltage is greater than at any of the error amplifiers (taking into account the built-in thresholds and offsets) - the circuit output is allowed. When the generator is reset from maximum to zero, the outputs are switched off. A trigger with paraphase output divides the frequency in half. With logical 0 at input 13 (output mode), the trigger phases are combined by OR and supplied simultaneously to both outputs; with logical 1, they are supplied in phase to each output separately.
Output transistors - npn Darlingtons with built-in thermal protection (but without current protection). Thus, the minimum voltage drop between the collector (usually closed to the positive bus) and the emitter (at the load) is 1.5 V (typical at 200 mA), and in a circuit with a common emitter it is a little better, 1.1 V typical. The maximum output current (with one open transistor) is limited to 500 mA, the maximum power for the entire chip is 1 W.
Switching power supplies are gradually replacing their traditional relatives in audio engineering, since they look noticeably more attractive both economically and in size. The same factor that switching power supplies contribute significantly to the distortion of the amplifier, namely the appearance of additional overtones, is no longer relevant mainly for two reasons - the modern element base makes it possible to design converters with a conversion frequency significantly higher than 40 kHz, therefore the power modulation introduced by the power supply will already be in ultrasound. In addition, a higher power supply frequency is much easier to filter, and the use of two L-shaped LC filters along the power supply circuits already sufficiently smoothes out the ripples at these frequencies.
Of course, there is a fly in the ointment in this barrel of honey - the difference in price between a typical power supply for a power amplifier and a pulsed one becomes more noticeable as the power of this unit increases, i.e. The more powerful the power supply, the more profitable it is in relation to its standard counterpart.
And that is not all. When using switching power supplies, it is necessary to adhere to the rules for installing high-frequency devices, namely the use of additional screens, feeding the power part of the common wire to the heat sinks, as well as correct ground wiring and connection of shielding braids and conductors.
After a short lyrical digression about the features of switching power supplies for power amplifiers, the actual circuit diagram of a 400W power supply:

Figure 1. Schematic diagram of a switching power supply for power amplifiers up to 400 W
ENLARGE IN GOOD QUALITY

The control controller in this power supply is TL494. Of course, there are more modern chips to perform this task, but we use this particular controller for two reasons - it is VERY easy to purchase. For quite a long time, TL494 from Texas Instruments was used in the manufactured power supplies; no quality problems were found. The error amplifier is covered by OOS, which makes it possible to achieve a fairly large coefficient. stabilization (ratio of resistors R4 and R6).
After the TL494 controller there is an IR2110 half-bridge driver, which actually controls the gates of the power transistors. The use of the driver made it possible to abandon the matching transformer, which is widely used in computer power supplies. The IR2110 driver is loaded onto the gates through the R24-VD4 and R25-VD5 chains that accelerate the closing of the field gates.
Power switches VT2 and VT3 operate on the primary winding of the power transformer. The midpoint required to obtain alternating voltage in the primary winding of the transformer is formed by elements R30-C26 and R31-C27.
A few words about the operating algorithm of the switching power supply on the TL494:
At the moment of supplying a mains voltage of 220 V, the capacitances of the primary power supply filters C15 and C16 are infected through resistors R8 and R11, which does not allow the diol bridge VD to be overloaded by a short circuit current of completely discharged C15 and C16. At the same time, capacitors C1, C3, C6, C19 are charged through a line of resistors R16, R18, R20 and R22, stabilizer 7815 and resistor R21.
As soon as the voltage on capacitor C6 reaches 12 V, the zener diode VD1 “breaks through” and current begins to flow through it, charging capacitor C18, and as soon as the positive terminal of this capacitor reaches a value sufficient to open thyristor VS2, it will open. This will turn on relay K1, which with its contacts will bypass current-limiting resistors R8 and R11. In addition, the opened thyristor VS2 will open transistor VT1 to both the TL494 controller and the IR2110 half-bridge driver. The controller will begin a soft start mode, the duration of which depends on the ratings of R7 and C13.
During a soft start, the duration of the pulses that open the power transistors increases gradually, thereby gradually charging the secondary power capacitors and limiting the current through the rectifier diodes. The duration increases until the secondary supply is sufficient to open the LED of optocoupler IC1. As soon as the brightness of the optocoupler LED becomes sufficient to open the transistor, the pulse duration will stop increasing (Figure 2).

Figure 2. Soft start mode.

It should be noted here that the duration of the soft start is limited, since the current passing through resistors R16, R18, R20, R22 is not enough to power the TL494 controller, the IR2110 driver and the switched-on relay winding - the supply voltage of these microcircuits will begin to decrease and will soon decrease to a value at which TL494 will stop generating control pulses. And it is precisely until this moment that the soft start mode must be completed and the converter must return to normal operation, since the TL494 controller and the IR2110 driver receive their main power from the power transformer (VD9, VD10 - midpoint rectifier, R23-C1-C3 - RC filter , IC3 is a 15 V stabilizer) and that is why capacitors C1, C3, C6, C19 have such large values ​​- they must maintain the controller’s power supply until it returns to normal operation.
The TL494 stabilizes the output voltage by changing the duration of control pulses of power transistors at a constant frequency - Pulse-Width Modulation - PWM. This is only possible if the value of the secondary voltage of the power transformer is higher than that required at the output of the stabilizer by at least 30%, but not more than 60%.

Figure 3. Operating principle of a PWM stabilizer.

As the load increases, the output voltage begins to decrease, the optocoupler LED IC1 begins to glow less, the optocoupler transistor closes, reducing the voltage on the error amplifier and thereby increasing the duration of the control pulses until the effective voltage reaches the stabilization value (Figure 3). As the load decreases, the voltage will begin to increase, the LED of optocoupler IC1 will begin to glow brighter, thereby opening the transistor and reducing the duration of the control pulses until the effective value of the output voltage decreases to a stabilized value. The amount of stabilized voltage is regulated by trimming resistor R26.
It should be noted that the TL494 controller does not regulate the duration of each pulse depending on the output voltage, but only the average value, i.e. the measuring part has some inertia. However, even with capacitors installed in the secondary power supply with a capacity of 2200 μF, power failures at peak short-term loads do not exceed 5%, which is quite acceptable for HI-FI class equipment. We usually install capacitors in the secondary power supply of 4700 uF, which gives a confident margin for peak values, and the use of a group stabilization choke allows us to control all 4 output power voltages.
This switching power supply is equipped with overload protection, the measuring element of which is the current transformer TV1. As soon as the current reaches a critical value, thyristor VS1 opens and bypasses the power supply to the final stage of the controller. The control pulses disappear and the power supply goes into standby mode, which it can remain in for quite a long time, since the thyristor VS2 continues to remain open - the current flowing through resistors R16, R18, R20 and R22 is enough to keep it in the open state. How to calculate a current transformer.
To exit the power supply from standby mode, you must press the SA3 button, which will bypass the thyristor VS2 with its contacts, the current will stop flowing through it and it will close. As soon as the contacts SA3 open, the transistor VT1 closes itself, removing power from the controller and driver. Thus, the control circuit will switch to minimum consumption mode - thyristor VS2 is closed, therefore relay K1 is turned off, transistor VT1 is closed, therefore the controller and driver are de-energized. Capacitors C1, C3, C6 and C19 begin to charge and as soon as the voltage reaches 12 V, the thyristor VS2 opens and the switching power supply starts.
If you need to put the power supply into standby mode, you can use the SA2 button, when pressed, the base and emitter of transistor VT1 will be connected. The transistor will close and de-energize the controller and driver. The control pulses will disappear, and the secondary voltages will disappear. However, the power will not be removed from relay K1 and the converter will not restart.
This circuit design allows you to assemble power supplies from 300-400 W to 2000 W, of course, some circuit elements will have to be replaced, since their parameters simply cannot withstand heavy loads.
When assembling more powerful options, you should pay attention to the capacitors of the primary power supply smoothing filters C15 and C16. The total capacitance of these capacitors must be proportional to the power of the power supply and correspond to the proportion 1 W of the output power of the voltage converter corresponds to 1 µF of the capacitance of the primary power filter capacitor. In other words, if the power of the power supply is 400 W, then 2 capacitors of 220 μF should be used, if the power is 1000 W, then 2 capacitors of 470 μF or two of 680 μF must be installed.
This requirement has two purposes. Firstly, the ripple of the primary supply voltage is reduced, which makes it easier to stabilize the output voltage. Secondly, using two capacitors instead of one facilitates the operation of the capacitor itself, since electrolytic capacitors of the TK series are much easier to obtain, and they are not entirely intended for use in high-frequency power supplies - the internal resistance is too high and at high frequencies these capacitors will heat up. Using two pieces, the internal resistance is reduced, and the resulting heating is divided between two capacitors.
When used as power transistors IRF740, IRF840, STP10NK60 and similar ones (for more information about the transistors most commonly used in network converters, see the table at the bottom of the page), diodes VD4 and VD5 can be abandoned altogether, and the values ​​of resistors R24 and R25 can be reduced to 22 Ohms - power The IR2110 driver is quite enough to control these transistors. If a more powerful switching power supply is being assembled, then more powerful transistors will be required. Attention should be paid to both the maximum current of the transistor and its dissipation power - switching stabilized power supplies are very sensitive to the correct installation of the snubber and without it, the power transistors heat up more because currents formed due to self-induction begin to flow through the diodes installed in the transistors. Read more about choosing a snubber.
Also, the closing time that increases without a snubber makes a significant contribution to heating - the transistor stays in linear mode longer.
Quite often they forget about one more feature of field-effect transistors - with increasing temperature, their maximum current decreases, and quite strongly. Based on this, when choosing power transistors for switching power supplies, you should have at least a two-fold maximum current reserve for power amplifier power supplies and a three-fold reserve for devices operating on a large, unchanging load, for example, an induction smelter or decorative lighting, powering low-voltage power tools.
The output voltage is stabilized using the group stabilization choke L1 (GLS). You should pay attention to the direction of the windings of this inductor. The number of turns must be proportional to the output voltages. Of course, there are formulas for calculating this winding unit, but experience has shown that the overall power of the core for a DGS should be 20-25% of the overall power of the power transformer. You can wind until the window is filled by about 2/3, not forgetting that if the output voltages are different, then the winding with a higher voltage should be proportionally larger, for example, you need two bipolar voltages, one at ±35 V, and the second to power the subwoofer with voltage ±50 V.
We wind the DGS into four wires at once until 2/3 of the window is filled, counting the turns. The diameter is calculated based on a current intensity of 3-4 A/mm2. Let's say we got 22 turns, let's make up the proportion:
22 turns / 35 V = X turns / 50 V.
X turns = 22 × 50 / 35 = 31.4 ≈ 31 turns
Next, I’ll cut two wires for ±35 V and wind up another 9 turns for a voltage of ±50.
ATTENTION! Remember that the quality of stabilization directly depends on how quickly the voltage changes to which the optocoupler diode is connected. To improve the stabilization coefficient, it makes sense to connect an additional load to each voltage in the form of 2 W resistors with a resistance of 3.3 kOhm. The load resistor connected to the voltage controlled by the optocoupler should be 1.7...2.2 times less.

The circuit data for network switching power supplies on ferrite rings with a permeability of 2000 Nm are summarized in Table 1.

However, it is not always possible to recognize the brand of ferrite, especially if it is ferrite from horizontal transformers of televisions. You can get out of the situation by finding out the number of turns experimentally. More details about this in the video:

Using the above circuitry of a switching power supply, several submodifications were developed and tested, designed to solve a particular problem at various powers. The printed circuit board drawings for these power supplies are shown below.
Printed circuit board for a switching stabilized power supply with a power of up to 1200...1500 W. Board size 269x130 mm. In fact, this is a more advanced version of the previous printed circuit board. It is distinguished by the presence of a group stabilization choke, which allows you to control the magnitude of all power voltages, as well as an additional LC filter. Has fan control and overload protection. The output voltages consist of two bipolar power sources and one bipolar low-current source, designed to power the preliminary stages.

External view of the printed circuit board for a power supply up to 1500 W. DOWNLOAD IN LAY FORMAT

A stabilized switching network power supply with a power of up to 1500...1800 W can be made on a printed circuit board measuring 272x100 mm. The power supply is designed for a power transformer made on K45 rings and located horizontally. It has two bipolar power sources, which can be combined into one source to power an amplifier with two-level power supply and one bipolar low-current source for preliminary stages.

Printed circuit board of a switching power supply up to 1800 W. DOWNLOAD IN LAY FORMAT

This power supply can be used to power high-power automotive equipment, such as powerful car amplifiers and car air conditioners. Board dimensions 188x123. The Schottky rectifier diodes used are parallelized by jumpers and the output current can reach 120 A at a voltage of 14 V. In addition, the power supply can produce bipolar voltage with a load capacity of up to 1 A (installed integrated voltage stabilizers no longer allow). The power transformer is made on K45 rings, the filtering power voltage choke is made on two K40x25x11 rings. Built-in overload protection.

External view of the printed circuit board of the power supply for automotive equipment DOWNLOAD IN LAY FORMAT

The power supply up to 2000 W is made on two boards measuring 275x99, located one above the other. The voltage is controlled by one voltage. Has overload protection. The file contains several options for the “second floor” for two bipolar voltages, for two unipolar voltages, for the voltages required for two and three level voltages. The power transformer is located horizontally and is made on K45 rings.

Appearance of a “two-story” power supply DOWNLOAD IN LAY FORMAT

A power supply with two bipolar voltages or one for a two-level amplifier is made on a board measuring 277x154. Has a group stabilization choke and overload protection. The power transformer is on K45 rings and is located horizontally. Power up to 2000 W.

External view of the printed circuit board DOWNLOAD IN LAY FORMAT

Almost the same power supply as above, but has one bipolar output voltage.

External view of the printed circuit board DOWNLOAD IN LAY FORMAT

The switching power supply has two power bipolar stabilized voltages and one bipolar low current. Equipped with fan control and overload protection. It has a group stabilization choke and additional LC filters. Power up to 2000...2400 W. The board has dimensions 278x146 mm

External view of the printed circuit board DOWNLOAD IN LAY FORMAT

The printed circuit board of a switching power supply for a power amplifier with two-level power supplies, measuring 284x184 mm, has a group stabilization choke and additional LC filters, overload protection and fan control. A distinctive feature is the use of discrete transistors to speed up the turn-off of power transistors. Power up to 2500...2800 W.

with two-level power supply DOWNLOAD IN LAY FORMAT

A slightly modified version of the previous PCB with two bipolar voltages. Size 285x172. Power up to 3000 W.

External view of the printed circuit board of the power supply for the amplifier DOWNLOAD IN LAY FORMAT

Bridged network switching power supply with a power of up to 4000...4500 W is made on a printed circuit board measuring 269x198 mm. It has two bipolar power voltages, fan control and overload protection. Uses group stabilization choke. It is advisable to use remote additional secondary power supply filters.

External view of the printed circuit board of the power supply for the amplifier DOWNLOAD IN LAY FORMAT

There is much more space for ferrites on boards than there could be. The fact is that it is not always necessary to go beyond the sound range. Therefore, additional areas are provided on the boards. Just in case, a small selection of reference data on power transistors and links to where I would buy them. By the way, I have ordered both TL494 and IR2110 more than once, and of course power transistors. It’s true that I didn’t take the entire assortment, but so far I haven’t come across any defects.

Most modern switching power supplies are made on chips like TL494, which is a pulse PWM controller. The power part is made from powerful elements, such as transistors. The connection circuit of the TL494 is simple, a minimum of additional radio components is required, it is described in detail in the datasheet.

Modification options: TL494CN, TL494CD, TL494IN, TL494C, TL494CI.

I also wrote reviews of other popular ICs.

• 1. Characteristics and functionality
• 2. Analogues
• 3. Typical connection diagrams for power supply on TL494
• 4. Power supply diagrams
• 5. Converting an ATX power supply into a laboratory one
• 6.Datasheet
• 7. Electrical characteristics graphs
• 8. Microcircuit functionality

Characteristics and functionality

The TL494 chip is designed as a PWM controller for switching power supplies, with a fixed operating frequency. To set the operating frequency, two additional external elements are required: a resistor and a capacitor. The microcircuit has a 5V reference voltage source, the error of which is 5%.

Scope of application specified by the manufacturer:

1. power supplies with a capacity of more than 90W AC-DC with PFC;
2. microwaves;
3. boost converters from 12V to 220V;
4. power supplies for servers;
5. inverters for solar panels;
6. electric bicycles and motorcycles;
7. buck converters;
8. smoke detectors;
9. desktop computers.

Analogues

The most famous analogues of the TL494 chip are the domestic KA7500B, KR1114EU4 from Fairchild, Sharp IR3M02, UA494, Fujitsu MB3759. The connection diagram is similar, the pinout may be different.

The new TL594 is an analogue of the TL494 with increased comparator accuracy. TL598 is an analogue of TL594 with a repeater at the output.

Typical connection diagrams for power supply on TL494

The basic circuits for switching on the TL494 are collected from datasheets from various manufacturers. They can serve as the basis for the development of similar devices with similar functionality.

Power supply circuits

I will not consider complex circuits of TL494 switching power supplies. They require a lot of parts and time, so making them yourself is not rational. It’s easier to buy a ready-made similar module from the Chinese for 300-500 rubles.

When assembling boost voltage converters, pay special attention to cooling the output power transistors. For 200W the output current will be about 1A, relatively not much. Testing for stability of operation should be carried out with the maximum permissible load. It is best to form the required load from 220 volt incandescent lamps with a power of 20w, 40w, 60w, 100w. Do not overheat the transistors by more than 100 degrees. Follow safety precautions when working with high voltage. Try it on seven times, turn it on once.

The boost converter on the TL494 requires virtually no adjustment and is highly repeatable. Before assembly, check the resistor and capacitor values. The smaller the deviation, the more stable the inverter will operate from 12 to 220 volts.

It is better to control the temperature of transistors using a thermocouple. If the radiator is too small, it is easier to install a fan so as not to install a new radiator.

I had to make a power supply for the TL494 with my own hands for a subwoofer amplifier in a car. At that time, 12V to 220V car inverters were not sold, and the Chinese did not have Aliexpress. As an amplifier, the ULF used an 80W TDA series microcircuit.

Over the past 5 years, interest in electrically driven technology has increased. This was facilitated by the Chinese, who began mass production of electric bicycles, modern wheel-motor with high efficiency. I consider two-wheeled and one-wheeled hoverboards to be the best implementation. In 2015, the Chinese company Ninebot bought the American Segway and began producing 50 types of Segway-type electric scooters.

A good control controller is required to control a powerful low voltage motor.

Converting an ATX power supply into a laboratory one

Every radio amateur has a powerful ATX power supply from a computer that produces 5V and 12V. Its power ranges from 200W to 500W. Knowing the parameters of the control controller, you can change the parameters of the ATX source. For example, increase the voltage from 12 to 30V. There are 2 popular methods, one from Italian radio amateurs.

Let's consider the Italian method, which is as simple as possible and does not require rewinding transformers. The ATX output is completely removed and modified according to the circuit. A huge number of radio amateurs have repeated this scheme due to its simplicity. Output voltage from 1V to 30V, current up to 10A.

Datasheet

The chip is so popular that it is produced by several manufacturers; offhand I found 5 different datasheets, from Motorola, Texas Instruments and other lesser known ones. The most complete datasheet TL494 is from Motorola, which I will publish.

All datasheets, you can download each one:

• Motorola;
• Texas Instruments - the best datasheet;
• Contek

[+] Supplemented with scale files and photographs.

Scheme and description of alterations

Rice. 1

Before upgrading, you need to check the UPS for functionality, otherwise nothing good will come of it.

Remove the 115/230V switch and sockets for connecting cords. In place of the upper socket we install a PA1 microammeter for 150 - 200 µA from cassette recorders, the original scale is removed, and a homemade scale made using the FrontDesigner program is installed instead, scale files are attached.

Using a metal cutter, we cut the tracks on the UPS printed circuit board going to pins No. 1, 2, 3, 4, 13, 14, 15, 16 of the DA1 microcircuit and solder the parts according to the diagram (Fig. 1).

We replace all electrolytic capacitors on the +12V bus with 25V capacitors. We connect the standard fan M1 through voltage regulator DA2.
During installation, it is also necessary to take into account that resistors R12 and R13 heat up during operation of the unit; they must be located closer to the fan.

Correctly assembled, without errors, the device starts up immediately. By changing the resistance of resistor R10, we check the limits of output voltage adjustment, approximately from 3 - 6 to 18 - 25 V (depending on the specific instance). We select a constant resistor in series with R10, limiting the upper limit of adjustment to the level we need (let’s say 14 V). We connect a load to the terminals (with a resistance of 2 - 3 Ohms) and by changing the resistance of resistor R4 we regulate the current in the load.

If +12 V 8 A was written on the UPS sticker, then you should not try to remove 15 Amperes from it.

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THE ARTICLE WAS PREPARED BASED ON THE BOOK BY A. V. GOLOVKOV and V. B LYUBITSKY “POWER SUPPLY SUPPLY FOR SYSTEM MODULES OF THE IBM PC-XT/AT TYPE” PUBLISHING HOUSE “LAD&N” Moscow 1995 downloaded in electronic form from the Internet

CONTROL IC TL494

In modern UPSs, specialized integrated circuits (ICs) are usually used to generate the control voltage for switching power transistors of the converter.
An ideal control IC to ensure normal operation of a UPS in PWM mode should satisfy most of the following conditions:
operating voltage not higher than 40V;
the presence of a highly stable thermally stabilized reference voltage source;
presence of a sawtooth voltage generator
providing the ability to synchronize a programmable soft start with an external signal;
the presence of a mismatch signal amplifier with high common-mode voltage;
presence of a PWM comparator;
presence of a pulse controlled trigger;
the presence of a two-channel pre-terminal cascade with short-circuit protection;
presence of double pulse suppression logic;
availability of means for correcting the symmetry of output voltages;
the presence of current limitation in a wide range of common-mode voltages, as well as current limitation in each period with shutdown in emergency mode;
availability of automatic control with direct transmission;
ensuring shutdown when the supply voltage drops;
providing surge protection;
ensuring compatibility with TTL/CMOS logic;
providing remote switching on and off.

Figure 11. TL494 control chip and its pinout.

In the vast majority of cases, a TL494CN type microcircuit manufactured by TEXAS INSTRUMENT (USA) is used as a control circuit for the class of UPS under consideration (Fig. 11). It implements most of the functions listed above and is produced by a number of foreign companies under different names. For example, the SHARP company (Japan) produces the IR3M02 microcircuit, the FAIRCHILD company (USA) - UA494, the SAMSUNG company (Korea) - KA7500, the FUJITSU company (Japan) - MB3759, etc. All these microcircuits are complete analogues of the domestic KR1114EU4 microcircuit. Let us consider in detail the design and operation of this control chip. It is specially designed to control the power part of the UPS and contains (Fig. 12):

Figure 12. Functional diagram of the TL494 IC

Ramp voltage generator DA6; the GPG frequency is determined by the values ​​of the resistor and capacitor connected to the 5th and 6th pins, and in the class of power supply under consideration is chosen to be approximately 60 kHz;
stabilized reference voltage source DA5 (Uref=+5,OB) with external output (pin 14);
dead zone comparator DA1;
comparator PWM DA2;
voltage error amplifier DA3;
error amplifier for current limit signal DA4;
two output transistors VT1 and VT2 with open collectors and emitters;
dynamic push-pull D-trigger in frequency division mode by 2 - DD2;
auxiliary logic elements DD1 (2-OR), DD3 (2ND), DD4 (2ND), DD5 (2-OR-NOT), DD6 (2-OR-NOT), DD7 (NOT);
constant voltage source with a rating of 0.1BDA7;
DC source with a nominal value of 0.7 mA DA8.
The control circuit will start, i.e. sequences of pulses will appear on pins 8 and 11 if any supply voltage is applied to pin 12, the level of which is in the range from +7 to +40 V. The entire set of functional units included in the TL494 IC can be divided into digital and the analog part (digital and analog signal paths). The analog part includes error amplifiers DA3, DA4, comparators DA1, DA2, sawtooth voltage generator DA6, as well as auxiliary sources DA5, DA7, DA8. All other elements, including output transistors, form the digital part (digital path).

Figure 13. Operation of the TL494 IC in nominal mode: U3, U4, U5 - voltages at pins 3, 4, 5.

Let us first consider the operation of the digital path. Timing diagrams explaining the operation of the microcircuit are shown in Fig. 13. From the timing diagrams it is clear that the moments of appearance of the output control pulses of the microcircuit, as well as their duration (diagrams 12 and 13) are determined by the state of the output of the logical element DD1 (diagram 5). The rest of the “logic” performs only the auxiliary function of dividing the output pulses of DD1 into two channels. In this case, the duration of the output pulses of the microcircuit is determined by the duration of the open state of its output transistors VT1, VT2. Since both of these transistors have open collectors and emitters, they can be connected in two ways. When switched on according to a circuit with a common emitter, the output pulses are removed from the external collector loads of the transistors (from pins 8 and 11 of the microcircuit), and the pulses themselves are directed downward from the positive level (the leading edges of the pulses are negative). The emitters of the transistors (pins 9 and 10 of the microcircuit) in this case are usually grounded. When switched on according to a circuit with a common collector, external loads are connected to the emitters of the transistors and the output pulses, directed in this case by surges (the leading edges of the pulses are positive), are removed from the emitters of transistors VT1, VT2. The collectors of these transistors are connected to the power bus of the control chip (Upom).
The output pulses of the remaining functional units that are part of the digital part of the TL494 microcircuit are directed upward, regardless of the circuit diagram of the microcircuit.
The DD2 trigger is a push-pull dynamic D flip-flop. The principle of its operation is as follows. On the leading (positive) edge of the output pulse of element DD1, the state of input D of flip-flop DD2 is written to the internal register. Physically, this means that the first of the two flip-flops included in DD2 is switched. When the pulse at the output of element DD1 ends, the second flip-flop within DD2 is switched along the falling (negative) edge of this pulse, and the state of the DD2 outputs changes (information read from input D appears at output Q). This eliminates the possibility of an unlocking pulse appearing at the base of each of the transistors VT1, VT2 twice during one period. Indeed, as long as the pulse level at input C of trigger DD2 has not changed, the state of its outputs will not change. Therefore, the pulse is transmitted to the output of the microcircuit through one of the channels, for example the upper one (DD3, DD5, VT1). When the pulse at input C ends, trigger DD2 switches, locks the upper channel and unlocks the lower channel (DD4, DD6, VT2). Therefore, the next pulse arriving at input C and inputs DD5, DD6 will be transmitted to the output of the microcircuit via the lower channel. Thus, each of the output pulses of element DD1, with its negative edge, switches trigger DD2 and thereby changes the channel of passage of the next pulse. Therefore, the reference material for the control microcircuit indicates that the architecture of the microcircuit provides double pulse suppression, i.e. eliminates the appearance of two unlocking pulses based on the same transistor per period.
Let us consider in detail one period of operation of the digital path of the microcircuit.
The appearance of an unlocking pulse based on the output transistor of the upper (VT1) or lower (VT2) channel is determined by the logic of the operation of elements DD5, DD6 (“2OR-NOT”) and the state of elements DD3, DD4 (“2AND”), which, in turn, , is determined by the state of trigger DD2.
The operating logic of the 2-OR-NOT element, as is known, is that a high-level voltage (logical 1) appears at the output of such an element in the only case where low voltage levels (logical 0) are present at both of its inputs. For other possible combinations of input signals, the output of element 2 OR-NOT has a low voltage level (logical 0). Therefore, if at the output Q of the trigger DD2 there is a logical 1 (moment ti of diagram 5 in Fig. 13), and at the output /Q there is a logical 0, then at both inputs of the element DD3 (2I) there will be logical 1 and, therefore, a logical 1 will appear at the output DD3, and therefore at one of the inputs of element DD5 (2OR-NOT) of the upper channel. Therefore, regardless of the level of the signal arriving at the second input of this element from the output of element DD1, the state of output DD5 will be logical O, and transistor VT1 will remain in the closed state. The output state of element DD4 will be logical 0, because logical 0 is present at one of the inputs of DD4, coming there from the /Q output of flip-flop DD2. Logical 0 from the output of element DD4 is supplied to one of the inputs of element DD6 and makes it possible for a pulse to pass through the lower channel. This pulse of positive polarity (logical 1) will appear at the output of DD6, and therefore at the base of VT2 during the pause between the output pulses of element DD1 (i.e. for the time when there is a logical 0 at the output of DD1 - interval trt2 of diagram 5, Fig. 13 ). Therefore, transistor VT2 opens and a pulse appears on its collector, ejecting it downward from the positive level (if connected according to a circuit with a common emitter).
The beginning of the next output pulse of element DD1 (moment t2 of diagram 5 in Fig. 13) will not change the state of the elements of the digital path of the microcircuit, with the exception of element DD6, at the output of which a logical 0 will appear, and therefore transistor VT2 will close. The completion of the output pulse DD1 (moment ta) will cause a change in the state of the outputs of the trigger DD2 to the opposite (logical 0 - at output Q, logical 1 - at output /Q). Therefore, the state of the outputs of elements DD3, DD4 will change (at the output of DD3 - logical 0, at the output of DD4 - logical 1). The pause that began at moment!3 at the output of element DD1 will make it possible to open transistor VT1 of the upper channel. Logical 0 at the output of element DD3 will “confirm” this possibility, turning it into the real appearance of an unlocking pulse based on transistor VT1. This impulse lasts until moment U, after which VT1 closes and the processes are repeated.
Thus, the main idea of ​​​​the operation of the digital path of the microcircuit is that the duration of the output pulse at pins 8 and 11 (or at pins 9 and 10) is determined by the duration of the pause between the output pulses of the DD1 element. Elements DD3, DD4 determine the channel for the passage of a pulse using a low-level signal, the appearance of which alternates at the outputs Q and /Q of the trigger DD2, controlled by the same element DD1. Elements DD5, DD6 are low level matching circuits.
To complete the description of the functionality of the microcircuit, one more important feature should be noted. As can be seen from the functional diagram in the figure, the inputs of elements DD3, DD4 are combined and output to pin 13 of the microcircuit. Therefore, if logical 1 is applied to pin 13, then elements DD3, DD4 will work as repeaters of information from outputs Q and /Q of trigger DD2. In this case, elements DD5, DD6 and transistors VT1, VT2 will switch with a phase shift of half a period, ensuring the operation of the power part of the UPS, built according to a push-pull half-bridge circuit. If logical 0 is applied to pin 13, then elements DD3, DD4 will be blocked, i.e. the state of the outputs of these elements will not change (constant logical 0). Therefore, the output pulses of element DD1 will affect elements DD5, DD6 in the same way. Elements DD5, DD6, and therefore the output transistors VT1, VT2, will switch without a phase shift (simultaneously). This mode of operation of the control microcircuit is used if the power part of the UPS is made according to a single-cycle circuit. In this case, the collectors and emitters of both output transistors of the microcircuit are combined for the purpose of increasing power.
The output voltage is used as a “hard” logical unit in push-pull circuits
internal source of the chip Uref (pin 13 of the chip is combined with pin 14).
Now let's look at the operation of the analog circuit of the microcircuit.
The state of the DD1 output is determined by the output signal of the PWM comparator DA2 (diagram 4), supplied to one of the DD1 inputs. The output signal of the comparator DA1 (Diagram 2), supplied to the second input of DD1, does not affect the state of the DD1 output in normal operation, which is determined by the wider output pulses of the PWM comparator DA2.
In addition, from the diagrams in Fig. 13 it is clear that when the voltage level changes at the non-inverting input of the PWM comparator (diagram 3), the width of the output pulses of the microcircuit (diagrams 12, 13) will change proportionally. In normal operation, the voltage level at the non-inverting input of the PWM comparator DA2 is determined only by the output voltage of the error amplifier DA3 (since it exceeds the output voltage of the DA4 amplifier), which depends on the level of the feedback signal at its non-inverting input (pin 1 of the microcircuit). Therefore, when a feedback signal is applied to pin 1 of the microcircuit, the width of the output control pulses will change in proportion to the change in the level of this feedback signal, which, in turn, changes in proportion to changes in the level of the UPS output voltage, because Feedback comes from there.
The time intervals between output pulses at pins 8 and 11 of the microcircuit, when both output transistors VT1 and VT2 are closed, are called “dead zones”.
Comparator DA1 is called a “dead zone” comparator, because it determines its minimum possible duration. Let's explain this in more detail.
From the timing diagrams in Fig. 13 it follows that if the width of the output pulses of the PWM comparator DA2 decreases for some reason, then starting from a certain width of these pulses, the output pulses of the comparator DA1 will become wider than the output pulses of the PWM comparator DA2 and begin to determine the output state of the logical element DD1, and therefore. width of the output pulses of the microcircuit. In other words, comparator DA1 limits the width of the output pulses of the microcircuit at a certain maximum level. The limitation level is determined by the potential at the non-inverting input of comparator DA1 (pin 4 of the microcircuit) in steady state. However, on the other hand, the potential at pin 4 will determine the range of width adjustment of the output pulses of the microcircuit. As the potential at pin 4 increases, this range narrows. The widest adjustment range is obtained when the potential at pin 4 is 0.
However, in this case there is a danger associated with the fact that the width of the “dead zone” may become equal to 0 (for example, in the case of a significant increase in the current consumed from the UPS). This means that the control pulses at pins 8 and 11 of the microcircuit will follow directly after each other. Therefore, a situation known as a “rack breakdown” may arise. It is explained by the inertia of the inverter’s power transistors, which cannot open and close instantly. Therefore, if you simultaneously apply a locking signal to the base of a previously opened transistor, and an unlocking signal to the base of a closed transistor (i.e., with a zero “dead zone”), then you will get a situation where one transistor has not yet closed, and the other is already open. Then a breakdown occurs along the transistor stand of the half-bridge, which consists in the flow of through current through both transistors. This current, as can be seen from the diagram in Fig. 5, bypasses the primary winding of the power transformer and is practically unlimited. Current protection does not work in this case, because current does not flow through the current sensor (not shown in the diagram; the design and principle of operation of the current sensors used will be discussed in detail in subsequent sections), which means that this sensor cannot output a signal to the control circuit. Therefore, the through current reaches a very large value in a very short period of time. This leads to a sharp increase in the power released on both power transistors and almost instantaneous failure (usually breakdown). In addition, the diodes of the power rectifier bridge can be damaged by an inrush of through current. This process ends with the blowing of the network fuse, which, due to its inertia, does not have time to protect the circuit elements, but only protects the primary network from overload.
Therefore the control voltage; supplied to the bases of power transistors must be formed in such a way that first one of these transistors is reliably closed, and only then the other is opened. In other words, between the control pulses supplied to the bases of the power transistors there must be a time shift that is not equal to zero (“dead zone”). The minimum permissible duration of the “dead zone” is determined by the inertia of the transistors used as power switches.
The architecture of the microcircuit allows you to adjust the minimum duration of the “dead zone” using the potential at pin 4 of the microcircuit. This potential is set using an external divider connected to the output voltage bus of the internal reference source of the Uref microcircuit.
Some UPS versions do not have such a divider. This means that after the soft start process is completed (see below), the potential at pin 4 of the microcircuit becomes equal to 0. In these cases, the minimum possible duration of the “dead zone” will still not become equal to 0, but will be determined by the internal voltage source DA7 (0, 1B), which is connected to the non-inverting input of the comparator DA1 with its positive pole, and to pin 4 of the microcircuit with its negative pole. Thus, thanks to the inclusion of this source, the width of the output pulse of the comparator DA1, and therefore the width of the “dead zone,” under no circumstances can become equal to 0, which means that “breakdown along the rack” will be fundamentally impossible. In other words, the architecture of the microcircuit includes a limitation on the maximum duration of its output pulse (the minimum duration of the “dead zone”). If there is a divider connected to pin 4 of the microcircuit, then after a soft start the potential of this pin is not equal to 0, therefore the width of the output pulses of the comparator DA1 is determined not only by the internal source DA7, but also by the residual (after the completion of the soft start process) potential at pin 4. However, at the same time, as mentioned above, the dynamic range of the width adjustment of the PWM comparator DA2 is narrowed.

STARTING DIAGRAM

The starting circuit is designed to obtain voltage that could be used to power the control microcircuit in order to start it after turning on the IVP to the supply network. Therefore, start-up means the startup of the control microcircuit first, without which the normal operation of the power section and the entire UPS circuit as a whole is impossible.
The starting circuit can be constructed in two different ways:
with self-excitation;
with forced stimulation.
A self-excited circuit is used, for example, in the GT-150W UPS (Fig. 14). The rectified network voltage Uep is supplied to the resistive divider R5, R3, R6, R4, which is the base for both power key transistors Q1, Q2. Therefore, through the transistors, under the influence of the total voltage on capacitors C5, C6 (Uep), a base current begins to flow through the circuit (+)C5 - R5 - R7 - 6-e Q1 - R6 - R8 - 6-e Q2 - the “common wire” of the primary side - (-)C6.
Both transistors are slightly opened by this current. As a result, currents of mutually opposite directions begin to flow through the collector-emitter sections of both transistors along the circuits:
through Q1: (+)C5 - +310 V bus - Q1 - 5-6 T1 -1-2 T2-C9- (-)C5.
through Q2: (+)C6 - C9 - 2-1 T2 - 6-5 T1 - Q2 - "common wire" of the primary side - (-)C6.

Figure 14. Self-excited startup diagram of the GT-150W UPS.

If both currents flowing through the additional (starting) turns 5-6 T1 in opposite directions were equal, then the resulting current would be 0, and the circuit would not be able to start.
However, due to the technological spread of the current amplification factors of transistors Q1, Q2, one of these currents is always greater than the other, because transistors are slightly open to varying degrees. Therefore, the resulting current through turns 5-6 T1 is not equal to 0 and has one direction or another. Let us assume that the current through transistor Q1 predominates (that is, Q1 is more open than Q2) and, therefore, the current flows in the direction from pin 5 to pin 6 of T1. Further reasoning is based on this assumption.
However, in fairness, it should be noted that the current through transistor Q2 may also be predominant, and then all the processes described below will relate to transistor Q2.
The flow of current through turns 5-6 of T1 causes the appearance of an EMF of mutual induction on all windings of the control transformer T1. In this case, (+) EMF occurs at pin 4 relative to pin 5 and an additional current flows into the base Q1 under the influence of this EMF, opening it slightly through the circuit: 4 T1 - D7-R9-R7-6-3 Q1 - 5 T1.
At the same time, (-) EMF appears at pin 7 of T1 relative to pin 8, i.e. the polarity of this EMF turns out to be blocking for Q2 and it closes. Next, positive feedback (POF) comes into play. Its effect is that as the current increases through the collector-emitter section Q1 and turns 5-6 T1, an increasing EMF acts on winding 4-5 T1, which, creating an additional base current for Q1, opens it to an even greater extent. This process develops like an avalanche (very quickly) and leads to the complete opening of Q1 and the locking of Q2. A linearly increasing current begins to flow through open Q1 and the primary winding 1-2 of the power pulse transformer T2, which causes the appearance of an EMF pulse of mutual induction on all windings of T2. An impulse from winding 7-5 T2 charges storage capacity C22. A voltage appears at C22, which is supplied as a supply to pin 12 of the TL494 type control chip IC1 and to the matching stage. The microcircuit starts up and generates rectangular pulse sequences at its pins 11, 8, with which the power switches Q1, Q2 begin to switch through the matching stage (Q3, Q4, T1). Pulse EMF of the nominal level appears on all windings of power transformer T2. In this case, the EMF from windings 3-5 and 7-5 constantly feeds C22, maintaining a constant voltage level on it (about +27V). In other words, the microcircuit begins to power itself through the feedback ring (self-feeding). The unit enters operating mode. The supply voltage of the microcircuit and the matching stage is auxiliary, acts only inside the block and is usually called Upom.
This circuit may have some variations, such as in the LPS-02-150XT switching power supply (made in Taiwan) for the Mazovia CM1914 computer (Fig. 15). In this circuit, the initial impetus for the development of the startup process is obtained using a separate half-wave rectifier D1, C7, which powers the resistive divider basic for power switches in the first positive half-cycle of the network. This speeds up the startup process, because... the initial unlocking of one of the keys occurs in parallel with the charging of high-capacity smoothing capacitors. Otherwise, the scheme works similarly to that discussed above.

Figure 15. Self-excited starting circuit in the LPS-02-150XT switching power supply

This scheme is used, for example, in the PS-200B UPS from LING YIN GROUP (Taiwan).
The primary winding of the special starting transformer T1 is switched on at half the mains voltage (at a nominal value of 220V) or at full voltage (at a nominal value of 110V). This is done for reasons so that the amplitude of the alternating voltage on the secondary winding T1 does not depend on the rating of the supply network. When the UPS is turned on, alternating current flows through the primary winding T1. Therefore, an alternating sinusoidal EMF with the frequency of the supply network is induced on the secondary winding 3-4 T1. The current flowing under the influence of this EMF is rectified by a special bridge circuit on diodes D3-D6 and smoothed out by capacitor C26. A constant voltage of about 10-11V is released at C26, which is supplied as a supply to pin 12 of the TL494 type control microcircuit U1 and to the matching stage. In parallel with this process, the capacitors of the anti-aliasing filter are charged. Therefore, by the time power is supplied to the microcircuit, the power stage is also energized. The microcircuit starts up and begins to generate sequences of rectangular pulses at its pins 8, 11, with which the power switches begin to switch through the matching stage. As a result, the block's output voltages appear. After entering the self-feeding mode, the microcircuit is powered from the +12V output voltage bus through the decoupling diode D8. Since this self-feeding voltage is slightly higher than the output voltage of the rectifier D3-D5, the diodes of this starting rectifier are locked, and it does not subsequently affect the operation of the circuit.
The need for feedback via diode D8 is optional. In some UPS circuits that use forced excitation, there is no such connection. The control microcircuit and the matching stage are powered from the output of the starting rectifier during the entire operating time. However, the ripple level on the Upom bus in this case is slightly higher than in the case of powering the microcircuit from the +12V output voltage bus.
To summarize the description of launch schemes, we can note the main features of their construction. In a self-excited circuit, the power transistors are initially switched, resulting in the appearance of a supply voltage for the Upom chip. In a circuit with forced excitation, Upom is first obtained, and as a result, power transistors are switched. In addition, in self-excited circuits, the Upom voltage is usually around +26V, and in forced-excited circuits, it is usually around +12V.
A circuit with forced excitation (with a separate transformer) is shown in Fig. 16.

Figure 16. Start-up circuit with forced excitation of the PS-200B switching power supply (LING YIN GROUP).

MATCHING CASCADE

A matching stage is used to match and decouple the high-power output stage from low-power control circuits.
Practical schemes for constructing a matching cascade in various UPSs can be divided into two main options:
transistor version, where external discrete transistors are used as switches;
transistorless version, where the output transistors of the control chip itself VT1, VT2 (in integrated version) are used as keys.
In addition, another feature by which matching stages can be classified is the method of controlling the power transistors of a half-bridge inverter. Based on this feature, all matching cascades can be divided into:
cascades with common control, where both power transistors are controlled using one common control transformer, which has one primary and two secondary windings;
cascades with separate control, where each of the power transistors is controlled using a separate transformer, i.e. There are two control transformers in the matching stage.
Based on both classifications, the matching cascade can be performed in one of four ways:
transistor with general control;
transistor with separate control;
transistorless with general control;
transistorless with separate control.
Transistor stages with separate control are rarely used or not used at all. The authors did not have the opportunity to encounter such an embodiment of the matching cascade. The remaining three options are more or less common.
In all variants, communication with the power stage is carried out using a transformer method.
In this case, the transformer performs two main functions: amplification of the control signal in terms of current (due to attenuation in voltage) and galvanic isolation. Galvanic isolation is necessary because the control chip and matching stage are on the secondary side, and the power stage is on the primary side of the UPS.
Let's consider the operation of each of the mentioned matching cascade options using specific examples.
In a transistor circuit with common control, a push-pull transformer pre-power amplifier on transistors Q3 and Q4 is used as a matching stage (Fig. 17).

Figure 17. Matching stage of the KYP-150W switching power supply (transistor circuit with common control).

Figure 18. Real shape of pulses on the collectors

The currents through diodes D7 and D9, flowing under the influence of the magnetic energy stored in the DT core, have the form of a decaying exponential. In the DT core, during the flow of currents through diodes D7 and D9, a changing (falling) magnetic flux acts, which causes the appearance of EMF pulses on its secondary windings.
Diode D8 eliminates the influence of the matching stage on the control chip through the common power bus.
Another type of transistor matching stage with general control is used in the ESAN ESP-1003R switching power supply (Fig. 19). The first feature of this option is that the output transistors VT1, VT2 of the microcircuit are included as emitter followers. Output signals are removed from pins 9 and 10 of the microcircuit. Resistors R17, R16 and R15, R14 are emitter loads of transistors VT1 and VT2, respectively. These same resistors form the basic dividers for transistors Q3, Q4, which operate in switch mode. Capacitances C13 and C12 are forcing and help speed up the switching processes of transistors Q3, Q4. The second characteristic feature of this cascade is that the primary winding of the control transformer DT has no output from the middle point and is connected between the collectors of transistors Q3, Q4. When the output transistor VT1 of the control chip opens, the divider R17, R16, which is the base for transistor Q3, is energized with voltage Upom. Therefore, current flows through control junction Q3 and it opens. The acceleration of this process is facilitated by the forcing capacitance C13, which supplies the Q3 base with an unlocking current that is 2-2.5 times higher than the established value. The result of opening Q3 is that the primary winding 1-2 DT is connected to the housing with its pin 1. Since the second transistor Q4 is locked, an increasing current begins to flow through the primary winding DT along the circuit: Upom - R11 - 2-1 DT - Q3 - housing.

Figure 19. Matching stage of switching power supply ESP-1003R ESAN ELECTRONIC CO., LTD (transistor circuit with common control).

Rectangular EMF pulses appear on the secondary windings 3-4 and 5-6 DT. The winding direction of the DT secondary windings is different. Therefore, one of the power transistors (not shown in the diagram) will receive an opening base pulse, and the other will receive a closing pulse. When VT1 of the control chip closes sharply, Q3 also closes sharply after it. The acceleration of the closing process is facilitated by the forcing capacitance C13, the voltage from which is applied to the base-emitter junction Q3 in the closing polarity. Then the “dead zone” lasts when both output transistors of the microcircuit are closed. Next, the output transistor VT2 opens, which means that the divider R15, R14, which is the base for the second transistor Q4, is powered by voltage Upom. Therefore, Q4 opens and the primary winding 1-2 DT is connected to the housing at its other end (pin 2), so an increasing current begins to flow through it in the opposite direction to the previous case along the circuit: Upom -R10- 1-2 DT - Q4 - "frame".
Therefore, the polarity of the pulses on the secondary windings of DT changes, and the second power transistor will receive the opening pulse, and a pulse of closing polarity will act on the basis of the first. When VT2 of the control chip closes sharply, Q4 also closes sharply after it (using the forcing capacitance C12). Then the “dead zone” continues again, after which the processes are repeated.
Thus, the main idea behind the operation of this cascade is that an alternating magnetic flux in the DT core can be obtained due to the fact that the primary winding DT is connected to the housing at one end or the other. Therefore, alternating current flows through it without a direct component with a unipolar supply.
In transistorless versions of the matching stages of the UPS, output transistors VT1, VT2 of the control microcircuit are used as transistors of the matching stage, as noted earlier. In this case, there are no discrete matching stage transistors.
A transistorless circuit with general control is used, for example, in the PS-200V UPS circuit. The output transistors of the microcircuit VT1, VT2 are loaded along the collectors by the primary half-windings of the transformer DT (Fig. 20). Power is supplied to the midpoint of the primary winding DT.

Figure 20. Matching stage of the PS-200B switching power supply (transistorless circuit with common control).

When transistor VT1 opens, an increasing current flows through this transistor and half-winding 1-2 of the control transformer DT. Control pulses appear on the secondary windings of DT, having such a polarity that one of the inverter power transistors opens and the other closes. At the end of the pulse, VT1 closes sharply, the current through half-winding 1-2 DT stops flowing, so the EMF on the secondary windings DT disappears, which leads to the closing of the power transistors. Next, the “dead zone” lasts when both output transistors VT1, VT2 of the microcircuit are closed, and no current flows through the primary winding DT. Next, transistor VT2 opens, and the current, increasing over time, flows through this transistor and half-winding 2-3 DT. The magnetic flux created by this current in the DT core has the opposite direction to the previous case. Therefore, an EMF of polarity opposite to the previous case is induced on the secondary windings DT. As a result, the second transistor of the half-bridge inverter opens, and at the base of the first, the pulse has a polarity that closes it. When VT2 of the control chip closes, the current through it and the primary winding DT stops. Therefore, the EMF on the secondary windings DT disappears, and the inverter power transistors are closed again. Then the “dead zone” continues again, after which the processes are repeated.
The main idea of ​​​​building this cascade is that an alternating magnetic flux in the core of the control transformer can be obtained by supplying power to the middle point of the primary winding of this transformer. Therefore, currents flow through the half-windings with the same number of turns in different directions. When both output transistors of the microcircuit are closed ("dead zones"), the magnetic flux in the core DT is equal to 0. Alternate opening of the transistors causes the alternate appearance of magnetic flux in one or the other half-winding. The resulting magnetic flux in the core is variable.
The last of these varieties (transistorless circuit with separate control) is used, for example, in the UPS of the Appis computer (Peru). In this circuit there are two control transformers DT1, DT2, the primary half-windings of which are collector loads for the output transistors of the microcircuit (Fig. 21). In this scheme, each of the two power switches is controlled through a separate transformer. Power is supplied to the collectors of the output transistors of the microcircuit from the common Upom bus through the midpoints of the primary windings of the control transformers DT1, DT2.
Diodes D9, D10 with the corresponding parts of the primary windings DT1, DT2 form core demagnetization circuits. Let's look at this issue in more detail.

Figure 21. Matching stage of the "Appis" switching power supply (transistorless circuit with separate control).

The matching stage (Fig. 21) is essentially two independent single-ended forward converters, because the opening current flows into the base of the power transistor during the open state of the matching transistor, i.e. the matching transistor and the power transistor connected to it through a transformer are open simultaneously. In this case, both pulse transformers DT1, DT2 operate with a constant component of the primary winding current, i.e. with forced magnetization. If special measures are not taken to demagnetize the cores, they will enter magnetic saturation over several periods of operation of the converter, which will lead to a significant decrease in the inductance of the primary windings and failure of the switching transistors VT1, VT2. Let's consider the processes occurring in the converter on transistor VT1 and transformer DT1. When transistor VT1 opens, a linearly increasing current flows through it and the primary winding 1-2 DT1 along the circuit: Upom -2-1 DT1 - circuit VT1 - “case”.
When the unlocking pulse at the base of VT1 ends, it closes abruptly. The current through winding 1-2 DT1 stops. However, the EMF on the demagnetizing winding 2-3 DT1 changes polarity, and the demagnetizing core DT1 current flows through this winding and diode D10 through the circuit: 2 DT1 - Upom - C9 - “body” - D10-3DT1.
This current is linearly decreasing, i.e. the derivative of the magnetic flux through the core DT1 changes sign, and the core is demagnetized. Thus, during this reverse cycle, the excess energy stored in the core DT1 during the open state of the transistor VT1 is returned to the source (storage capacitor C9 of the Upom bus is recharged).
However, this option for implementing the matching cascade is the least preferable, because both transformers DT1, DT2 operate with underutilization in induction and with a constant component of the primary winding current. The magnetization reversal of cores DT1, DT2 occurs in a private cycle, covering only positive induction values. Because of this, magnetic fluxes in the cores turn out to be pulsating, i.e. contain a constant component. This leads to increased weight and size parameters of transformers DT1, DT2 and, in addition, compared to other matching cascade options, two transformers are required here instead of one.